FIG. 1 is a block diagram showing prior art filter circuitry as described in Japanese patent application publication (TOKKAIHEI) No. 10-126212 (referred to as reference 1 from here on), for example. In the figure, reference numeral 1 denotes a SAW resonator that constitutes a parallel element, reference numeral 2 denotes a SAW resonator that constitutes a series element, reference numeral 3 denotes an input terminal, reference numeral 4 denotes a ground terminal disposed on an input side of the prior art filter circuitry, reference numeral 5 denotes an output terminal, and reference numeral 6 denotes a ground terminal disposed on an output side of the prior art filter circuitry.
FIG. 2 is a block diagram showing each SAW resonator. For example, in the case of the SAW resonator 1 that is a parallel element as shown in FIG. 1, the SAW resonator 1 has an upper electric terminal 7 with the same potential as the input terminal 3, and a lower electric terminal 8 with the same potential as the ground terminal 4 disposed on the input side of the filter circuitry, as shown in FIG. 2.
FIG. 3 is a drawing showing a concrete structure of the SAW resonator 1. In the figure, reference numeral 9 denotes an IDT (Inter Digital Transducer). A number of electrode fingers 10 each having a width of d1 are alternately arranged at predetermined spacings of P1 over a length W. Each electrode finger 10 is typically composed of a metallic thin film having a thickness of h, which is mainly composed of aluminum or which can be mainly composed of a metallic material other than aluminum. Reference numeral 11 denotes a reflector in which a number of metallic strips 12 each having a width of d2 are arranged at predetermined spacings of P2, as in the case of the IDT 9.
Each reflector 11 as shown in FIG. 3 is a short-circuited strip reflector in which all metallic strips 12 are connected to one another so that they have the same potential. As an alternative, each reflector 11 can be an open-circuited strip in which all metallic strips 12 are not connected to one another so that they have their respective potentials. The spacings between the IDT 9 and the two reflectors 11 are g1 and g2 that can be equal to each other in most cases.
When an electrical signal is applied between the electric terminals 7 and 8, an electric field appears between two electrode fingers 10 that are adjacent to each other and results in the excitation of a surface acoustic wave. At this time, the surface acoustic wave is efficiently excited when the spacing P1 of the plurality of electrode fingers 10 is equal to one-half of the wavelength λ of the surface acoustic wave. In other words, the spacing P1 of the plurality of electrode fingers 10 determines the operating frequency of the SAW resonator. When a normal piezoelectric substrate is used, the surface acoustic wave excited between any two adjacent electrode fingers 10 propagates in two directions perpendicular to the plurality of electrode fingers 10, i.e., in two directions respectively extending from the IDT 9 to the two reflectors 11.
On the other hand, in each reflector 11, the differences among the mass loads and electric boundary conditions of the plurality of metallic strips 12 can cause a reflection of the surface acoustic wave at edges of the plurality of metallic strips 12. When the spacing P2 of the plurality of metallic strips 12 is equal to the one-half wavelength λ/2 of the surface acoustic wave, a strong reflection is caused because all reflected waves at the edges of the plurality of metallic strips 12 are in phase with one another.
In this case, the surface acoustic wave excited by the IDT 9 can reflect from the two reflectors 11 located on the both sides of the IDT 9, so that the energy of the surface acoustic wave is confined in the SAW resonator and the SAW resonator thus serves as a resonator. The operation of the SAW resonator is disclosed in detail by “Acoustic wave device technological handbook”, 1st ed., pp. 217 to 227 (referred to as reference 2 from here on), which was edited by Japan Society for the Promotion of Science 150th Committee on Acoustic Wave Device Technology and which was published on Nov. 30, 1991). The SAW resonator has an input impedance that is minimized at a resonance frequency fr, and an input admittance that is minimized at an antiresonance frequency fa. In addition, the resonance frequency fr is lower than the antiresonance frequency fa.
FIG. 4 is a circuit diagram showing an equivalent circuit of the SAW resonator. In the figure, reference numeral 13 denotes an electrode capacitance C0 that the IDT 9 of FIG. 3 has, reference numeral 14 denotes an inductor L1, and reference numeral 15 denotes a capacitor C1.
The resonance frequency fr of the SAW resonator is equal to the frequency of a series resonance circuit comprised of the inductor 14 and the capacitor 15, and the impedance between the electric terminals 7 and 8 of the SAW resonator therefore becomes nearly-short-circuited at the resonance frequency fr.
Furthermore, the antiresonance frequency fa of the SAW resonator is equal to the frequency of a parallel resonance circuit comprised of the electrode capacitance 13 and the series circuit (i.e., the combination of the inductor 14 and the capacitor 15), and the impedance between the electric terminals 7 and 8 of the SAW resonator therefore becomes nearly-open-circuited at the antiresonance frequency fa. These relationships are given by the following equations:
                              f          r                =                  1                      2            ⁢                                                  ⁢            π            ⁢                                                            L                  1                                ⁢                                  C                  1                                                                                        (        1        )                                          f          a                =                                            1                              2                ⁢                                                                  ⁢                π                                      ⁢                                                            1                                                            L                      1                                        ⁢                                          C                      1                                                                      ⁢                                  (                                                                                    C                        1                                                                    C                        0                                                              +                    1                                    )                                                              =                                    f              r                        ⁢                                          1                +                                                      C                    1                                                        C                    0                                                                                                          (        2        )            
The reference 2 discloses an equivalent circuit in which the inductor 14 is assumed to have a resistance component R1 and a q-factor (Quality Factor) is taken into account when series resonance occurs. In the case of such an equivalent circuit, the impedance between the electric terminals 7 and 8 of the SAW resonator at the resonance frequency fr does not become completely-short-circuited, but has a minimum value.
FIG. 5 is an explanatory drawing for explaining an operation of the prior art filter circuitry of FIG. 1. FIG. 5A shows the impedance characteristics of the SAW resonator 2 that is a series element, FIG. 5B shows the admittance characteristics of the SAW resonator 1 that is a parallel element, and FIG. 5 C shows the filter characteristics of the prior art filter circuitry in which the SAW resonator 2 that is a series element and the two SAW resonators 1 each of which is a parallel element are connected as shown in FIG. 1.
Next, a description will be made as to the operation of the prior art filter circuitry.
The SAW resonator 2 that is the series element produces series resonance at a frequency fr2, and produces parallel resonance at a frequency fra2. In other words, the resonance frequency of the SAW resonator 2 is fr2 and the antiresonance frequency of the SAW resonator 2 is fa2.
The vertical axis of FIG. 5A shows an imaginary part of the impedance of the SAW resonator 2. The SAW resonator 2 serves as a capacitor having a capacitance of C0 in a frequency range in which no surface acoustic wave is excited. Therefore, the SAW resonator 2 has a negative imaginary impedance in a range of frequencies lower from the resonance frequency fr2 and in a range of frequencies higher than the antiresonance frequency fa2.
On the other hand, each of the two SAW resonators 1 that is a parallel element produces series resonance at a frequency fr1, and produces parallel resonance at a frequency fa1. In other words, the resonance frequency of the SAW resonator 1 is fr1 and the antiresonance frequency of the SAW resonator 1 is fa1.
The vertical axis of FIG. 5B shows an imaginary part of the admittance of each SAW resonator 1. The imaginary part of the admittance of each SAW resonator 1 has a positive admittance in a range of frequencies lower from the resonance frequency fr1 and in a range of frequencies higher than the antiresonance frequency fa1.
Here, the resonance frequency fr2 of the SAW resonator 2 is set so that it is nearly equal to the antiresonance frequency fa1 of each SAW resonator 1. At this time, because the SAW resonator 2 has an impedance that is nearly equal to 0 at a frequency which is close to the resonance frequency fr2, the SAW resonator 2 becomes a short-circuited state. On the other hand, because each of the two SAW resonators 1 has an admittance that is nearly equal to 0 at a frequency which is close to the antiresonance frequency fr1, each SAW resonator 1 becomes a nearly-open-circuited state. Therefore, a nearly-short-circuit occurs between the input terminal 3 and the output terminal 5, and a nearly-open-circuit occurs between the input terminal 3 and the ground terminal 4 disposed on the input side of the filter circuitry and a nearly-open-circuit similarly occurs between the output terminal 5 and the ground terminal 6 disposed on the input side of the filter circuitry. The filter circuitry disposed between the input terminal 3 and the output terminal 5 thus has a low-loss passband.
On the other hand, each SAW resonator 1 becomes a nearly-short-circuited state at a frequency that is close to the resonance frequency fr1. In this case, because a nearly-short-circuit occurs between the input terminal 3 and the ground terminal 4 disposed on the input side of the filter circuitry and a nearly-short-circuit also occurs between the output terminal 5 and the ground terminal 6 disposed on the output side of the filter circuitry, no electrical signal can be transmitted from the input terminal 3 to the output terminal 5 and a large attenuation pole is formed. Because this attenuation pole has a frequency that is close to the resonance frequency fr1 of the SAW resonator 1, the frequency of the attenuation pole is limited to a frequency that is lower than the antiresonance frequency fa1 of the SAW resonator 1 that is placed in the passband of the filter circuitry.
The SAW resonator 2 becomes a nearly-open-circuited state at a frequency that is close to the antiresonance frequency fa2. In this case, no electrical signal can be transmitted from the input terminal 3 to the output terminal 5 and a large attenuation pole is formed. Because this attenuation pole has a frequency that is close to the antiresonance frequency fa2 of the SAW resonator 2, the frequency of the attenuation pole is limited to a frequency that is higher than the resonance frequency fr2 of the SAW resonator 2 that is placed in the passband of the filter circuitry.
Even when using resonators other than the SAW resonators as the series element and two parallel elements included in the filter circuitry, the filter circuitry of FIG. 1 exhibits similar characteristics. For example, the filter circuitry of FIG. 1 exhibits similar characteristics even when using bulk wave resonators that utilize a thickness longitudinal vibration or thickness slip vibration as the series element and two parallel elements included in the filter circuitry.
For example, as disclosed in “Basic of solid vibration theory for electric and electronics”, 1st ed., pp. 175 to 188 (referred to as reference 3 from here on), which was published on Sep. 20, 1982 by Ohmsha Co. and which was supervised by Morio Onoe, it is known that a bulk wave resonator has the following approximately-established relationship among a resonance frequency fr, an antiresonance frequency fa, and an electromechanical coupling constant k2 of a piezoelectric element that constitutes the bulk wave resonator.k2˜2(fa−fr)/fa  (3)
This equation (3) shows that the difference between the resonance frequency fr and antiresonance frequency fa of the bulk wave resonator is nearly equal to one-half of the electromechanical coupling constant k2 of the used piezoelectric element which is multiplied by the antiresonance frequency fa. This relationship is similarly established for SAW resonators. In other words, when filter circuitry is comprised of acoustic wave resonators such as bulk wave resonators or SAW resonators, because the difference between the center of frequencies within the passband of the filter circuitry and the frequency of an attenuation pole that provides large attenuation is equal to the difference between the resonance frequency fr and antiresonance frequency fa of an acoustic wave resonator included in the filter circuitry, the difference between the resonance frequency fr and antiresonance frequency fa of the acoustic wave resonator is limited to almost one-half of the center of frequencies within the passband of the filter circuitry which is multiplied by the electromechanical coupling constant k2 of the used piezoelectric element. Therefore, the difference between the center of the passband of the filter circuitry and the center of the stopband of the filter circuitry that needs large attenuation is limited by the performance of the used piezoelectric element.
For example, although either lithium niobate (LiNb03) or lithium tantalate (LiTa03) is widely known and used as the piezoelectric element that is used in each SAW resonator included in the filter circuitry, such a piezoelectric element has an electromechanical coupling constant k2 of at most ten and a few %. A problem is therefore that the difference between the center of the passband of the filter circuitry and the frequency of an attenuation pole that provides large attenuation is only about 5 to 6% of the center of the passband of the filter circuitry.
FIG. 6 is a block diagram showing other prior art filter circuitry as disclosed in Japanese patent application publication (TOKKAIHEI) No. 6-350390 (referred to as reference 4 from here on), for example. The other prior art filter circuitry is so constructed as to include, as a series element, a series circuit comprised of a first SAW resonator 2a, a second SAW resonator 2b, and an inductor 16, and, as a parallel element, a parallel resonance circuit comprised of an inductor 15 and a capacitor 14.
FIG. 7 is an explanatory drawing for explaining an operation of the other prior art filter circuitry of FIG. 6. FIG. 7A shows an impedance characteristic 17 of the first SAW resonator 2a, an impedance characteristic 18 of the second SAW resonator 2b, and an impedance characteristic 19 of the inductor 16.
The first SAW resonator 2a has a resonance frequency of fr1 and an antiresonance frequency of fa1, and the second SAW resonator 2b has a resonance frequency of fr2 and an antiresonance frequency of fa2. The antiresonance frequency fa1 of the first SAW resonator 2a is lower than the resonance frequency fr2 of the second SAW resonator 2b. 
FIG. 7B shows an admittance characteristic 20 of the parallel resonance circuit comprised of the inductor 14 and the capacitor 15. The parallel resonance circuit has an antiresonance frequency of fap, and the antiresonance frequency fap of the parallel resonance circuit is so set as to be intermediate between the antiresonance frequency fa1 of the first SAW resonator 2a and the resonance frequency fr2 of the second SAW resonator 2b. 
The vertical axis of each of FIGS. 7A and 7B shows an imaginary part. FIG. 7C shows a pass filter characteristic of the filter circuitry constructed as show in FIG. 6.
Because the antiresonance frequency fap of the parallel resonance circuit that consists of the inductor 14 and the capacitor 15 is set to a frequency fpass that falls within the passband of the filter circuit, the parallel resonance circuit has an impedance that becomes nearly-open-circuited. Because the first SAW resonator 2a operates at a frequency higher than the antiresonance frequency fa1 thereof, the first SAW resonator 2a has a capacitive impedance. In addition, because the second SAW resonator 2b operates at a frequency higher than the resonance frequency fr2 thereof, the second SAW resonator 2b has a capacitive impedance. Therefore, the inductor 16 having an inductivity impedance, is indispensable in order to cancel out the capacitive impedances of the first and second SAW resonators 2a and 2b. In general, an inductor has a large loss at a frequency that is of the order of GHz. For example, an inductor formed on a dielectric substrate has a q-factor of the order of several tens, and a high-q-factor inductor, such as an air-core coil, has a q-factor of at most the order of 100. Therefore, in the case where an inductor is included in each of the series element and parallel element of the filter circuitry as shown in FIG. 6, there is a problem that loss in the passband increases.
Furthermore, in the parallel resonance circuit that is the parallel element, the admittance of the inductor 14 becomes predominant at a frequency lower than the antiresonance frequency fap of the parallel resonance circuit because it is smaller than the admittance of the capacitor 15, and the parallel resonance circuit exhibits an inductive admittance. On the other hand, the admittance of the capacitor 15 becomes predominant at a frequency higher than the antiresonance frequency fap of the parallel resonance circuit because it is smaller than the admittance of the inductor 14, and the parallel resonance circuit exhibits a capacitive admittance. A further problem encountered with the prior art filter circuitry is therefore that because at a frequency within the passband the filter circuitry has an impedance having a component other than a pure resistance component, which increases with increase in the difference between the frequency and the antiresonance frequency fap, it is difficult to implement low-loss characteristics over a wide frequency band.
FIG. 8 is a block diagram showing prior art filter circuitry as shown in Japanese patent application publication No. (TOKKAIHEI) No. 9-116380 (referred to as reference 5 from here on), for example. In the figure, reference numeral 21 denotes a resonance circuit in which a capacitor 13 and a series resonance circuit (i.e., a capacitor 15 and an inductor 14) are connected in parallel with each other.
The resonance circuit 21 is the same as the one as shown in FIG. 4, and, essentially, it is no different from a SAW resonator. In addition, from the viewpoint of design, the resonance frequency of the resonance circuit 21 is so set as to be nearly equal to the antiresonance frequency of each SAW resonator 1 that is a parallel element, as in the case of the prior art filter circuitry as shown in FIG. 1.
While the difference between the resonance frequency and antiresonance frequency of a SAW resonator is determined dependent upon the electromechanical coupling constant of a piezoelectric element used in the SAW resonator, the resonance circuit 21 as shown in FIG. 8 does not have such a limitation and therefore the filter circuitry including the resonance circuit 21 has room for broadening of the passband. However, the q-factor of the inductor included in the resonance circuit 21 is considerably smaller than the q-factor of the SAW resonator 2 of the filter circuitry of FIG. 1. Another problem is therefore that while a wider-band characteristic can be implemented as compared with the filter circuitry including the SAW resonator 2, it is difficult to implement a low-loss pass filter characteristic.
A further problem is that because the q-factor of the resonance circuit 21 is small, it is difficult for the prior art filter circuitry to provide steeper attenuation at frequencies higher than the passband of the filter circuit, which is produced by the series element of the filter circuitry and it is difficult for the attenuation pole which the resonance circuit 21 forms to produce a steep zero point, so that the attenuation characteristic degrades in a frequency band higher than the passband of the filter circuitry.
In other words, a problem with prior art filter circuitry constructed as mentioned above is that when frequencies within the passband are away from those within the attenuation band, it is difficult to implement a low-loss wide-band pass filter characteristic and it is also difficult to provide a large attenuation over a wide frequency band.
The present invention is proposed to solve the above-mentioned problem, and it is therefore an object of the present invention to provide filter circuitry that can implement a low-loss wide-band pass filter characteristic and can provide a large attenuation over a wide frequency band, even though frequencies within the passband are away from those within the attenuation band.